ElectroOptical Innovationshttps://electrooptical.net/News/2022-02-02T18:22:07+00:00Signal to Noise Ratio and You, Part 12021-01-24T11:16:17+00:002022-02-02T18:22:07+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/digital-lock-in-principles/<p>In building an ultrasensitive instrument, we're always fighting to improve our signal-to-noise ratio (SNR). The SNR is the ratio of signal power to noise power in the measurement bandwidth, and is limited by noise in the instrument itself and the noise of any background signals, such as the shot noise of the background light or the slight hiss of a microphone. </p>
<p>If the signal is weak, it will have proportionally more noise, so that the apparatus has to be designed to get rid of as much noise as possible. There are a number of ways to do this. The best is to get more signal or reduce the noise, for instance by increasing the laser power and using a <a href="https://www.electrooptical.net/Projects/laser-noise-cancellers/">laser noise canceller</a>, but eventually we hit a practical limit. At that point, we're left with several options, all of which boil down to filtering in one form or another.</p>
<p>Filters can be hardware or software, but their job is to pass the desired signal frequencies and reject noise at other frequencies. Of course some of the noise lands on top of our signal and so makes it through the filter anyway.</p>
<p>A low-pass filter passes frequencies below its cutoff and attenuates higher ones. If the signal is concentrated below the cutoff frequency, the filter rejects the high-frequency noise while preserving the signal (and the low-frequency noise, of course). By slowing down the measurement, for example by reducing the scan speed, the bandwidth of the signal's frequency spectrum can be reduced and the filter made correspondingly narrower.</p>
<p> A problem with this simple approach is that in most cases there's a concentration of noise at low frequencies (near DC), so filtering doesn't help as much as one might expect--in fact, it's not uncommon for the noise to get <em>worse</em> as the measurement gets slower, which is rather unintuitive. It's because there is a lower limit to the signal spectrum as well as an upper. If we're taking 1000 measurements, each with an averaging time of a millisecond, then the signal spectrum is predominantly contained between 1 Hz and 1 kHz. A measurement that takes a second doesn't contain much signal information or noise between 0 Hz (DC) and 1 Hz. Slowing it down to one measurement per hundred seconds reduces the lower cutoff to (1/100) Hz and the upper cutoff to 10 Hz. That narrows the bandwidth, all right, but interestingly it typically makes the noise worse rather than better. Let's look at why.</p>
<p>To find the total noise, we have to add up the noise contributions at all frequencies in the filter passband. In other words, the total noise power is the integral of the noise power spectral density (PSD). The low frequency noise PSD often goes like 1/<em>f</em>, whose integral is ln(<em>f</em>). Thus if the passband is between <em>f</em><sub>1</sub> and <em>f</em><sub>2</sub>, the total noise goes as ln(<em>f</em><sub>2</sub>) - ln(<em>f</em><sub>1</sub>) = ln(<em>f</em><sub>2 </sub>/ <em>f</em><sub>1</sub>). Because the ratio <em>f</em><sub>2 </sub>/ <em>f</em><sub>1</sub> is the same in both the fast and slow measurements, the 1/<em>f</em> noise is also the same—sacrificing a factor of 100 in speed hasn't improved things at all. In fact, since things like thermal drifts rise more steeply than 1/<em>f</em>, going slower is likely to make things worse in real cases. So lowpass filtering can help, but only up to a point. In <a href="https://electrooptical.net/News/signal-to-noise-ratio-and-you-part-2/">Part 2</a>, we'll look at ways to get round this roadblock.</p>
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<p>A slightly more sophisticated approach that generally works better is to make the signal periodic in time at some frequency <em>f</em>, i.e. to move it away from DC to escape the low-frequency noise.&nbsp; (This is generally easy to do.)&nbsp; Our noise rejection filter now needs to be a bandpass centered at f, and we'll also need some means of measuring the amplitude and phase of the AC signal.&nbsp; That's more complicated, of course, but with that setup we can narrow the bandwidth as much as we like and still get the full SNR improvement.&nbsp; A lock-in amplifier is a device for making such narrow-band AC measurements conveniently.</p>
<p>An AC signal that passes through a narrow filter becomes a sine wave with some amplitude and phase: <em>g(t) = A</em> cos(2<em>ft</em> + &phi;), where the signal information is contained in slowish variations of A and &phi;, the amplitude and phase. This is familiar from radio: you can send music and speech over the air by encoding it as amplitude modulation (AM) or angle modulation. Amplitude modulation changes the size of the peaks of the sinusoidal <em>carrier</em> wave in response to the audio signal (<em>A</em> varies), while angle modulation changes the position of the peaks in time (&phi; varies). There are two common types of intentional angle modulation: the more familiar frequency modulation (FM) or phase modulation (PM), which differ only in the details of how the signal amplitude is mapped onto the carrier phase. Both types of modulation widen the carrier spectrum, forming sidebands above and below <em>f</em> that carry the signal information. It's generally preferable to talk about phase modulation, especially in discussions of noise, because in PM a flat baseband noise spectrum produces flat sidebands, whereas in FM it doesn't. From elementary trigonometry, we know that cos(<em>a+b</em>) = cos a cos b - sin a sin b, or in this case, cos(2 pi f t + &phi;) = cos(2 pi f t) cos &phi; - sin(2 pi f t) sin &phi;. Thus by measuring the amplitudes of the sine and cosine components of the signal, we can recover its amplitude and phase. Rearranging the same trigonometric identity shows us how to do this: cos a cos b = ( cos(a-b) + cos(a+b) ) / 2 and sin a sin b = ( cos(a-b) - cos(a+b) ) / 2. Thus if we multiply our signal by _local oscillator_ (LO) signals sin(2 &pi; f t) and cos(2 &amp;pi. f t), we get I = A cos phi [cos(2 pi f t) ][cos( 2 pi f t)] = A cos phi [cos(0) + cos(4 pi f t)], which is I = A cos &phi; + (a signal near 2f), and Q = A sin &phi; [sin( 2 &pi; f t) ] [sin( 2 &pi; f t )] = A sin &phi; [ cos(0) cos(4 &pi; f t) ], which is Q = A sin &phi; + (another signal near 2f). Lowpass filtering gets rid of the 2f components of I and Q and rejects noise exactly as our narrow bandpass filter would, with the same tradeoff of bandwidth vs. measurement speed but without the excess low-frequency noise. I and Q are the so-called in-phase and quadrature signals, which are concentrated near DC in the so-called _baseband. You can think of "quadrature" as referring to the signal shifted a quarter cycle. Thus the procedure of multiplying by the sine and cosine phases of the carrier converts the modulated carrier into a pair of baseband signals containing both the amplitude and phase information. Because of the lowpass filtering, the exact waveform of the modulated wave (sine, square, or something else) doesn't matter much--only the sinusoidal components sufficiently close to f contribute. This property of sines and cosines is called _orthogonality_. Very often only one of the two is of interest, usually I, but one can also recover A and &phi; easily: A = sqrt( I2 + Q2 ) and &phi; = atan(Q/I). (One has to worry about a few other things when computing &phi;, such as which quadrant it's in, whether you're dividing by zero, and whether it needs unwrapping to avoid ambiguities of multiples of 2 &pi;.) The multiplications also of course produce the cross terms, proportional to cos(2 pi f t) sin(2 &pi; f t) = sin(4 &pi; f t)/2 but these have no baseband component and so get filtered out as well, showing that the sine and cosine components are orthogonal even though their frequencies are the same. A lock-in is basically a radio that measures the phase and amplitude of its input using two multipliers, one for I and one for Q, with the sine and cosine LO signals derived from a reference frequency that you supply. Generally this reference is the same source used to generate the AC modulation of the measured signal. There are two basic kinds of lock-ins: analog, where the multipliers and filters are physical circuits, and digital, where the signal is first digitized and the multipliers and narrow filters are done numerically by software or programmable logic. Either way, the orthogonality of sines and cosines is what makes lock-ins work. A fine but important point is that accurate digitization requires that the signal first pass through an analog filter to prevent high frequency junk from appearing at lower frequencies, a phenomenon called _aliasing_. This is familiar from moire' patterns in bridge railings and fences seen from the highway, or the tendency of stagecoach wheels in old Western movies to appear to rotate slowly backwards instead of quickly forwards. If the digitizer is sampling at f_s samples per second, the antialiasing filter has to reject frequencies above f_s/2, the so-called _Nyquist frequency_. (This requirement follows from the sampling theorem.) Real-world antialiasing filters are not infinitely sharp, so they have to start rolling of sooner than that. The maximum useful signal frequency is thus a bit below Nyquist, typically by 20%-30%. . Lock-ins are of course amplifiers as well; the amplification is mostly done ahead of the multipliers, and is generally range-switched rather than continuously variable like a volume control. It's a lot easier to make the amplifier quiet and stable that way, and those things matter a great deal in a lock-in. Because the signal of interest is often very much smaller than the wideband noise, lock-ins have to have a lot of _dynamic reserve_. Dynamic reserve is the ratio of the maximum allowable (signal + noise) to the full-scale signal amplitude on a given range, and is often a factor of 100 to 10,000 (40 to 80 decibels). The smallness of the desired signal is why the amplifiers have to be so stable and quiet, and the multipliers and the digitizer as well. (Minor problems in the digitizer system become very objectionable for this reason--in my experience nobody gets their first digital lock-in design quite right, because they aren't paranoid enough about this.) This is more than compensated for by the massive increase in multiplication accuracy and stability afforded by computer arithmetic compared with analog multiplier chips. Thus if it's done properly, a digital lock-in is better than an analog one, other things being equal. Done badly, it can easily be much worse. Quantization Noise ~~~~~~~~~~~~~~ Digital lock-ins also exhibit quantization noise, which requires a bit of explanation. An M-bit digitizer measuring a voltage V produces an M-bit binary fraction F = V/Vref, where V_ref is the reference voltage supplied to the digitizer. Digitizers come in various resolutions, usually between 10 and 24 bits. A plot of the output code vs. input voltage thus looks like a staircase, ideally a perfectly straight staircase with perfectly equal tread widths. The analog section of the digitizer contributes noise like any normal circuit, but in addition the digitizing operation introduces _quantization noise_, the inaccuracy inherent in converting a continuously-variable voltage into one of those 2M discrete steps. This is inherently a complicated thing to model, but we're saved by Widrow's theorem, which says that as long as the signal is at least a few steps in amplitude, the digitizing operation can be accurately modelled by a noiseless digitizer acting on a signal with added uniformly-distributed (white) noise of amplitude N = V<sub>ref&lt;\sub&gt; 2-M / &amp;surd;12. Mathematically the digital signal behaves just like a slightly noisier version of the analog one. Interestingly, the wideband noise provides an important benefit by exercising a much wider range of digitizer steps than the signal alone, effectively smoothing out minor irregularities in the staircase. (Pseudorandom noise is sometimes added in analog and subtracted again digitally to ensure that this happens in a known way, a procedure called _dithering_.) Sampling Rate Because the antialiasing filter is not adjustable, the digitizer must be run at a sufficiently-fast f<sub>s</sub> that the out-of-band components and higher harmonics of the modulation are attenuated enough that they don't reduce the accuracy of the measurement. There is thus little advantage in adjusting f<sub>s</sub>, so it's generally fixed in a given instrument. That means that even near the upper signal frequency limit the digitizer samples more than twice per cycle of the input signal, and at lower frequency many more times than that. Digitizers, Averaging, and Widrow's Theorem ~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~ Lock-ins use adjustable adjustable lowpass filters on I and Q to allow the user to trade off noise rejection vs. measurement speed. These lowpass filters form an average of the sampled values of I and Q. Averaging P samples of the signal reduces the filter bandwidth to f_s/(2P) and reduces the noise amplitude by a factor of 1/sqrt(P), so using narrower filters will reduce noise but require a slower measurement. Traditionally, lock-ins have used 1- or 2-pole RC analog filters, which work very similarly to simple digital filters used for continuous averaging of sampled data; the stored average (analog or digital) undergoes a slow exponential decay with time and new information is added to replace it. In this way, choosing a 1-s time constant results in the output being a moving exponential average of the past 4 or 5 seconds' worth of data. At lower carrier frequencies, the The Stanford Research Systems SRS 850 Digital Lock-In Amplifier The SRS 850 works in the above fashion, with a few additional details. It uses an 18-bit digitizer, a fixed sampling frequency of 256 kHz, and an antialiasing filter cutoff of 108 kHz. It allows a maximum reference frequency of 102.4 kHz . Its digital filters can be adjusted for time constants from 10 us to 30 ks (8.33 hours). It forms the LO signals by computing sines and cosines digitally to an accuracy of 24 bits, the word size of its internal digital signal processor. This is the same relative precision as an IEEE-standard 32-bit floating-point number, which has a 24-bit significand (23 bits plus one sign bit) and an 8-bit exponent. To do this, it must advance the numerical phase by 2 pi f_ref/f_s per sample. The reference frequency is continally measured and the numerical phase step and phase offset adjusted so that the positive-going zero-crossing of the cosine LO coincides with the positive-going zero-crossing of the reference. This is an example of a _digital phaselocked loop_ (DPLL).</sub></p>
-->Mirror of www.analog-innovations.com (Jim Thompson's site)2020-07-14T07:56:59+00:002022-01-20T14:36:29+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/mirror-of-wwwanalog-innovationscom-jim-thompsons-site/<p>James Elbert (Jim) Thompson was a well-known chip designer who used to be a regular on sci.electronics.design. He last posted in July 2018. As he was very sick at the time, we presume that he has died, but no obituary has so far turned up. He was born on February 29th, 1940, and used to say that he was looking forward to his 21st birthday in 2024.</p>
<p>Jim had a consulting company, Analog Innovations LLC, and a website, <a href="https://web.archive.org/web/20180808224712/http://analog-innovations.com/" target="_blank">http://www.analog-innovations.com</a>. You can find it on web.archive.org, but those copies are incomplete. A complete archive (minus a bit of javascript) is here at</p>
<p><a href="https://electrooptical.net/static/oldsite/www.analog-innovations.com/logo.html" target="_blank">https://electrooptical.net/static/oldsite/www.analog-innovations.com/logo.html</a></p>
<p>(main frame) and</p>
<p><a href="https://electrooptical.net/static/oldsite/www.analog-innovations.com/analog-innovations.html" target="_blank">https://electrooptical.net/static/oldsite/www.analog-innovations.com/analog-innovations.html</a></p>
<p>(navigation panel).</p>
<p>Jim designed many highly influential ICs, some of whose datasheets are posted on his site.</p>
<p>May God hold him in memory eternal.</p>Low Frequency Noise In InGaAs Heterojunction FETs2020-02-23T09:01:14+00:002022-01-20T11:36:28+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/low-frequency-noise-in-ingaas-heterojunction-fets/<p>InGaAs heterojunction FETs are magic parts—fast, strong, and extremely quiet. They're also called pseudomorphic high electron-mobility transistors (pHEMTs), because they use a 2D quantum well to to force the conduction electrons to move in a plane without much scattering. My fave Avago ATF38143 pHEMT was discontinued, but luckily Mini-Circuits stepped into the breach with their very nice <a href="https://www.minicircuits.com/pdfs/SAV-551+.pdf">SAV-551+</a> and its siblings, which are similar enough that the ATF SPICE model can be hacked up to work with them. (RF companies like Mini-Circuits never seem to supply SPICE models for some reason.) In one post on the 'purpose of precision' thread on sci.electronics.design, I noted that the Avago ATF38143 model I had <a href="http://web.archive.org/web/20200302203525/http://www.edaboard.co.uk/phil-t562709.html">posted awhile back</a> predicted way, way too much low frequency noise. The real pHEMTs tend to have a pretty accurately 1/<em>f</em> PSD with corner frequencies between 10 and 50 MHz and flatband noise of around 0.3 nV/√Hz, about 10 dB quieter than the best JFETs, as well as being 20 times faster.</p>
<p>Their 1/<em>f</em> characteristic and very high transconductance makes them pretty good for bootstraps. Capacitive transducers such as photodiodes have admittances that rise with frequency, because <em>Xc</em> = 1/(2π<em>f C</em>). With an amplifier whose noise is flat, this produces a power spectral density (PSD) proportional to <em>f</em><sup>2</sup>. (There's more on this in the docs for the <em><a href="https://hobbs-eo.com/QL01">QL01 from Hobbs ElectroOptics</a></em> , my old <a href="https://www.electrooptical.net/www/frontends/frontends.pdf">photodiode front ends paper</a>, or in Chapter 18 of <a href="https://electrooptical.net/Building_ElectroOptical_Systems/">Building ElectroOptical Systems</a>.)</p>
<p>This extra noise gets a bit ugly at high frequency, but here it has the nice side effect of suppressing the low frequency<em></em> noise: that factor of <em>f</em><sup>2</sup> squashes the 1/<em>f</em> noise pretty thoroughly, so these pHEMTs can be used at low frequencies with no issues. Their high transconductance makes them low output impedance followers, which is just what you need for a bootstrap. (Of course they can oscillate at frequencies as high as 12 GHz, so you do have to pay close attention to stability and good layout.)</p>
<p>Here's the old one:</p>
<p>.MODEL ATF38143_chip NMF( vto=-0.75, Beta=0.3, Lambda=0.07, Alpha=4,<br/>+ B=0.8, Pb=0.7, Cgs=0.997E-12,<br/>+ Cgd=0.176E-12, Rd=0.084, Rs=0.054, Kf=1e6, Af=1)</p>
<p>The relevant parameters are AF, which is the reciprocal of the noise exponent, and KF, which is the noise amplitude.</p>
<p>This produces a low-frequency noise power spectral density (PSD) that goes as 1/<em>f</em><sup>2</sup>, and the noise at 10 MHz is 340 <em>millivolts</em>/sqrt(Hz), where with a 20-MHz corner frequency, it should be around 500 <em>picovolts</em>/sqrt(Hz). The PSD goes as the voltage squared, so at 10 MHz the old model is pessimistic by 20 log(0.34 V/500 pV) = 177 dB!</p>
<p>Here's the new one, with AF = 2, which gives a 1/<em>f</em> PSD, and KF = 5E-11, which gives a 1/f corner of 22 MHz. (VTO is also tweaked to make it an enhancement device.)</p>
<p>.MODEL SAV551_chip NMF( vto=0.08, Beta=0.3, <br/>+ Lambda=0.07, Alpha=4 B=0.8, Pb=0.7, <br/>+ Cgs=0.997E-12, Cgd=0.176E-12, Rd=0.084, <br/>+ Rs=0.054, Kf=5e-11, Af=2)</p>
<p>Click on the image for the LTspice source file, sav551noise.asc.</p>
<p><a href="https://electrooptical.net/www/sed/sav551noise.asc"><img alt="Screen shot of schematic and noise plot" src="https://electrooptical.net/static/media/uploads/sed/sav551noise.png" width="95%"/></a></p>
<p>Cheers</p>
<p>Phil Hobbs</p>High-Value Ceramic Capacitors: They Stink, and You Can't Get Them Anyway2018-07-26T14:10:44+00:002022-01-20T06:02:21+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/high-value-ceramic-capacitors-they-stink-and-you-cant-get-them-anyway/<p>There are widespread shortages of electronics parts at the moment, especially passives. Quoted factory lead times are 40 weeks or thereabouts, and since the industry is capacity-limited, it isn't clear that the situation is going to get better any time soon, so everybody's starting to panic. Given all this churn I've been spending an unconscionable amount of time lately finding suitable replacements for out-of-stock parts. </p>
<p>High value ceramic caps are the worst--their capacitance drops by at least 60% and at worst 95% at rated voltage, so finding an adequate substitute involves a lot more than the package, value and voltage rating. Most of their data sheets are useless, which is frustrating. However, all is not lost: most makers have websites where you can look at the C(V) curves. </p>
<p>Here's an alphabetical list. Many of these links can also be used for resistors, inductors, and other component types as well.<br/><br/>AVX <a href="https://spicat.avx.com">SpiCat</a> (This one is super clunky--every time you select something it displays a throbber for 5-10 seconds. It does give you soakage models though, which is nice if they're vaguely accurate.)<br/><br/>Cornell Dubilier has a <a href="http://www.cde.com/technical-support/life-temperature-calculators">lifetime vs temperature-calculator</a><br/><br/>Kemet <a href="http://ksim.kemet.com">KSIM</a> (About the best; slow on some browsers)<br/><br/>CAD for <a href="https://global.kyocera.com/cgi-bin/electro/cap_download/cdas.cgi?LANG=EN">Kyocera capacitors</a><br/><br/>Murata <a href="http://ds.murata.co.jp/simsurfing/mlcc.html?lcid=en-us">SimSurfing</a> (Honourable mention):<br/><br/>Panasonic has downloadable <a href="https://industrial.panasonic.com/ww/design-support/ds-tools">selection tools</a> that don't run under WINE, so I don't know if they're any good.<br/><br/>Taiyo Yuden <a href="https://ds.yuden.co.jp/TYCOMPAS/or/searcherMain">TY Compas</a> (Honourable mention):<br/><br/>TDK <a href="https://product.tdk.com/info/en/index.html">tools</a><br/><br/>Yageo used to have selection tools at www dot yageo dot com/portal/product/product dot jsp but they've apparently gone away<br/><br/>Samsung has <a href="http://weblib.samsungsem.com/mlcc/mlcc-ec.do">some good searchable datasheets</a> that you can get to from e.g. Digikey's product page, but a lot of their products are the pits, e.g. <a href="https://pdf.datasheet.live/datasheets-1/samsung_electro-mechanics/CL31A106KBHNNNE.pdf">this one</a>, whose capacitance falls off by over 90% at rated voltage. Note that you want the <em>characteristics</em> link and not the <em>datasheet</em> link.<em> </em><br/><br/>Others don't seem to, e.g. Johanson, Vishay, and most of the smaller Chinese outfits. Sure would be nice if everybody had decent datasheets like Samsung's better ones.</p>Temperature Measurement is Hard2018-03-09T09:52:10+00:002020-07-16T06:40:48+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/temperature-measurement-is-hard/<p>Measuring temperature is surprisingly subtle. There are lots of sensors out there; Digikey sells thermistor sensors interchangeable to +- 0.1 C from several vendors for about $3 in onesies. IC sensors tout good accuracy and linearity, and come in both analogue and digital versions for way under a buck. So what's the issue?</p>
<p></p>
<p>The issue is: temperature sensors measure the temperature of the <em>sensor</em>, whereas what we want is the temperature of something else: air, fluid, or some solid object we're trying to control. So the problem is to get the sensor temperature to track the temperature we actually care about. IC sensors are especially bad, because they have stout leads made of copper (400 W/m/K thermal conductivity) and small packages made of plastic (0.1 W/m/K). Thus they basically measure the temperature of their leads, and are horrible at measuring air temperature, for instance.</p>
<p></p>
<p>National Semiconductor used to put out a very useful Temperature Measurements Handbook. Since TI bought them, it seems to have disappeared from the web, so here's the <a href="https://electrooptical.net/www/appnotes/NationalSemiTemperatureMeasurementHandbook2007.pdf">2007 edition.</a> There are lots of newer IC temperature sensors now, but not much has changed about the properties of plastic and metal, so the discussion in the first few pages about different sensor types and measurement difficulties is still very current.</p>Decap Photo of Terabeam APD Photoreceiver2018-02-24T20:53:26+00:002022-01-20T13:21:29+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/decap-photo-of-terabeam-apd-photoreceiver/<p><a href="https://electrooptical.net/static/oldsite/www/sed/TerabeamCD3109_decap2.jpg">Decap picture</a> of a Terabeam CD3109 APD/TIA module, taken with a lens glued to a cell<br/>phone camera</p>Thermal Runaway Found Useful2018-02-24T20:51:39+00:002022-01-20T12:06:28+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/thermal-runaway-found-useful/<p><a href="https://electrooptical.net/static/oldsite/www/sed/TemperatureBalancer.png">This odd circuit </a> is an <em>on-chip temperature balancer</em> that uses thermal runaway to force N transistor arrays to all run at the same temperature. BJT dissipation goes up at low temperature, with very high gain. Here's its <a href="https://electrooptical.net/static/oldsite/www/sed/TemperatureBalancerSteppResp100usPerDIv.tif">step response.</a><br/><br/></p>Sine Wave Generation with TANH Wave Shaper2018-02-24T20:46:04+00:002022-01-20T12:51:29+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/sine-wave-generation-with-tanh-wave-shaper/<p>Sine wave generation is a perennial problem.</p>
<p><br/>Direct-digital synthesis (DDS) uses a bunch of counters, lookup tables, and DACs,<br/>but that's a relatively heavyweight solution that doesn't fit all problems.<br/>BJT differential pairs naturally have a hyperbolic tangent (tanh) characteristic, which can be used to round off a triangle wave into a very passable sine. I'm not old enough to have invented this technique, but here are a couple of illustrations of how it works: <a href="https://electrooptical.net/static/oldsite/www/sed//TanhSineWaveShaper.pdf">TANH Sine Wave Shaper (PDF) </a>and <a href="https://electrooptical.net/static/oldsite/www/sed//TanhSineWaveShaper.mcd">TANH Sine Wave Shaper (Mathcad)</a>.<br/><br/></p>Care and Feeding of Tantalum Capacitors2018-02-24T20:40:06+00:002022-01-20T13:26:16+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/care-and-feeding-of-tantalum-capacitors/<p>Solid tantalum capacitors have a lot of advantages: very low inductance in surface mount packages, ESR low but not so low that your LDO regulators start oscillating; and good capacitance per unit volume. Unfortunately they're also prone to burn up when mistreated, which makes many engineers wary of them. This war story, entitled <a href="https://electrooptical.net/static/oldsite/www/sed/TantalumCapReforming_25272-what_a_cap_astrophe.pdf">What a Cap-astrophe! </a>talks about how to treat them properly, and how a bit of TLC after soldering can restore their full performance.<br/><br/></p>High Dynamic Range FET Bridge Mixers2018-02-24T20:34:12+00:002022-01-20T12:28:27+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/high-dynamic-range-fet-bridge-mixers/<p>For RF folks, one of the perennial quests is for better frequency mixers: lower distortion,<br/>lower power, better spurious performance. FETs can help. Nowadays CMOS muxes are the devices of choice for HF mixers, but to get the best performance, you still have to know how they work. Ed Oxner was a long-time Siliconix apps guy, and his paper on <a href="https://electrooptical.net/static/oldsite/www/sed/EdOxnerHighDynamicRangeFET_Mixers1985.pdf">High dynamic range FET RF mixers</a> is still right up there with the best. <br/>(From a Siliconix databook, 1985.) The FET mux approach is often credited to Dan Tayloe, but since they work just the same, the "Tayloe mixer" should really be called the "Oxner mixer".</p>Noise Peaks From Linear Voltage Regulators2018-02-24T15:10:29+00:002022-01-20T06:31:13+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/noise-peaks-from-linear-voltage-regulators/<p>Erroll Dietz is a remarkable fellow. He started out at National Semiconductor as Bob Pease's<br/>technician, and rose to become Chief Technology Officer.</p>
<p><br/>Feedback amplifiers generally have an output impedance that rises linearly with frequency---in other words, it's<br/>inductive. As <a href="https://electrooptical.net/static/oldsite/www/sed/ErrolDietzRegulatorNoisePeaks.pdf">Dietz's short paper</a> from Electronic Design shows, this effective inductance can<br/>resonate with the output bypass capacitor to cause really nasty noise peaks in the 1-100 kHz region. If you have an inexplicable noise peak in that range, a small resistor (a few tenths of an ohm to an ohm or two) in series with the regulator output can be just the ticket. You can also put it in series with the cap, but if you do that, make sure it's a pulse-withstanding type, or it's liable to blow up by an output short-circuit, or even the inrush transient, e.g. if somebody runs your gizmo off batteries. </p>
<p>Modern high-value multilayer ceramic caps (MLCCs) and aluminum-polymer electrolytics can have very low effective series resistance (ESR). This makes them amazing for filtering, but when used with voltage regulators, there's a catch: they make these noise peaks much worse, and frequently cause instability: the regulator's feedback loop goes into oscillation. This is really bad news; not only does your gizmo not work right, but the peaks of the oscillation are often way out of regulation, and so may destroy parts of the circuit. Low dropout (LDO) regulators, negative linear regulators, and most switchers are especially vulnerable to this.</p>
<p>Parallelling the Al-poly or MLCC filter with a tantalum or aluminum electrolytic of about the same value will often fix the problem, or you can use the pulse-withstanding resistor trick.</p>Random resources from my Usenet posts on sci.electronics.design2018-02-24T15:03:20+00:002020-01-18T01:31:18+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/random-resources-from-my-sed-posts/<p>Sometimes it's useful to add supporting documents, schematics, scope photos, simulations or other things to Usenet posts. By popular request I've added links to some of these, in no particular order.</p>
<p>A trick for turning the usual proportional/integral (PI) loop filter of a phaselocked loop (PLL) into a <a href="https://electrooptical.net/static/oldsite/sed/PLLSweeper2.asc"> sweep generator</a> for lock acquisition.</p>
<p>A big slide deck with ideas on <a href="https://electrooptical.net/static/oldsite/talks/GettingPDRight11.pdf">Getting Photodetection Right </a>(from a talk given at BBN in Cambridge MA)</p>
<p></p>
<p>A presentation from Texas Instruments showing detailed measurements of the <a href="https://electrooptical.net/static/oldsite/sed/OPA2188ChopperNoise.pdf">switching noise of their OPA2188 chopper amp</a>.</p>
<p>A paper from LIGO on <a href="https://electrooptical.net/static/oldsite/papers/LIGO_FlickerNoiseOfComponentsPaper.PDF">components with low flicker noise</a></p>How To Read SED: Archive Sites and Newsfeeds2018-02-24T14:34:57+00:002019-12-24T15:58:24+00:00Philip Hobbshttps://electrooptical.net/News/author/pcdh/https://electrooptical.net/News/how-to-read-sed-archive-sites-and-newsfeeds/<p>One of the nice things about sci.electronics.design is that it's widely redistributed by archive sites, some of which you can also use for posting, which is good since they made such a hash of the new <a href="https://groups.google.com/d/forum/sci.electronics.design">Google Groups</a> one.<br/><br/>Some examples:<br/><a href="https://www.electronics-related.com/groups/sci.electronics.design/1.php">electronics-related</a></p>
<p><a href="https://sci.electronics.design.narkive.com/">Narkive</a></p>
<p><a href="https://www.electrondepot.com/electrodesign/">Electron Depot</a></p>
<p>It's really better to use a proper newsreader such as Thunderbird or Forte Agent. You can get a free Usenet account from <a href="https://www.eternal-september.org">Eternal September</a> or <a href="https://www.aioe.org">aioe.org</a>.</p>Cascode Enhancement pHEMT Photodiode Preamp2017-09-07T12:30:22+00:002017-09-07T21:25:38+00:00Simon Hobbshttps://electrooptical.net/News/author/simon/https://electrooptical.net/News/cascode-enhancement-phemt-photodiode-preamp/<p>This is pretty small, because it has to be--those are microwave transistors, and will oscillate at the slightest provocation. The axial resistors and TO-92 parts are all for biasing---the actual amplifier is the part between the output coupling cap (the small orange thing in the middle) and the photodiode, which is the white square with the black middle at the right.</p>
<p><br/> The 0.1-inch pitch holes round the outside and 25-mil pad pitch set the scale.</p>
<p><br/> This amp works fine below 6 mA of bias current, but above there it wants to oscillate at 6 GHz. Interestingly this is just what SPICE predicts if the capacitance across the cascode device's base isolation bead is about 0,2 pF, which is a plausible number.</p>Transistor Tester for laser noise canceller2017-09-07T12:27:50+00:002018-02-24T20:14:50+00:00Simon Hobbshttps://electrooptical.net/News/author/simon/https://electrooptical.net/News/transistor-tester-for-laser-noise-canceller/<p>As discussed in medium-gory detail in <a href="https://www.electrooptical.net/static/media/uploads/sed/withouttears.pdf">this paper</a>, laser noise cancellers can let you do shot-noise limited measurements at baseband with lasers that are as much as 70 dB noisier than that.</p>
<p>They're limited by two main effects: beta nonlinearity (1/<i>h<sub>FE</sub></i>-1/<i>h<sub>fe</sub></i>) and log nonconformance (d ln(<i>I<sub>C</sub></i>)/d<i>V<sub>BE</sub></i> - <i>kT/e</i>).</p>
<p>This tester measures both of these quantities directly. It's a one-off, of course, so it's done with discrete logic and instrumentation amp parts. It has certain points of interest, for instance the use of a unity gain instrumentation amplifier as a precision +1/+2 gain amplifier. One loose end: U1 is a LT1043 switched-capacitor building block—a glorified MUX that has very low charge injection and very good common-mode rejection.</p>
<h6><a href="https://www.electrooptical.net/static/media/uploads/sed/noisecancellertransistortesterjuly6_2013.pdf">Schematic and data for the transistor tester</a></h6>